1. Field of the Invention
The present invention relates to radio frequency (RF) amplifiers. More particularly, the present invention relates to a multi-tone amplifier and a method for amplifying having adaptive closed-loop control for minimizing intermodulation distortion products, and for minimizing amplifier performance degradation caused by component drift, temperature variation and aging.
2. Description of the Related Art
As is well-known, when a dual or multi-tone input signal is applied to an amplifier that is not perfectly linear, undesirable intermodulation (IM) products are generated at predictable frequencies causing intermodulation distortion (IMD). Amplifiers operating in class AB or class B modes tend to produce high levels of IMD product when multi-frequency signals--that is, multi-tone signals--are amplified. IM product levels on the order of -30 dBc (30 decibels below the fundamental frequency or carrier level) are typical. The undesirable IM products are particularly apparent when the amplifier is operated in saturation or in the gain compression region of the amplifier. The level of the IM products are greater the further into the gain compression region the amplifier is operated.
Harmonic IM products are not of primary concern because they can be removed by a filter. Third order and fifth order IM products, however, fall within the desired communication bandwidth of the amplifier and cannot be removed by a filter. The only way to deal with the third and fifth order intermodulation products is to amplify in a way that does not generate third and fifth order intermodulation products.
A conventional technique for reducing intermodulation distortion (IMD) is to use a correction amplifier that generates correction signals at the same frequencies as the undesirable intermodulation (IM) products, but having phases that are 180.degree. out-of-phase from the phases of the corresponding IM products. When the IM products and the correction signals are applied to an output combiner, the IM products are cancelled by vector summation with the correction signals. As a result, the amplified output signal has substantially only the fundamental input signal frequencies, i.e., the multi-tone components of the input signal.
FIG. 1 shows a schematic block diagram of a conventional low-distortion RF amplifier circuit 10 that includes a correction amplifier. Circuit 10 linearly amplifies an input signal S.sub.IN to produce an amplified output signal S.sub.OUT. Input signal S.sub.IN is a dual-tone high-frequency signal having sinusoidal components at a first fundamental frequency f1 and at a second fundamental frequency f2. For this description, frequency f2 is greater than frequency f1. Both frequencies f1 and f2 are within standard wireless communication frequency bands, such as between 800-960 MHz. The phase of S.sub.IN is arbitrary. Amplification causes IM products to occur at both higher and lower frequencies than the communication frequency band of interest.
In FIG. 1, frequency components f1 and f2 of S.sub.IN and various other signals are shown vectorally for conveniently showing phase relationships between the same frequency components at specific points within circuit 10. Power and voltage standing wave ratio (VSWR) losses are ignored in the following description.
Input signal S.sub.IN is applied to an input port 11 of a first coupler, or power splitter, C1. Coupler C1 splits signal S.sub.IN into a signal S1 that is output at a "direct path" output port 12 and a signal S2 that is output at a "coupled path" output port 13. Typically, coupler C1 is a passive device, such as conventional branch line coupler or Wilkinson-type divider, that splits input power unequally between output ports 12 and 13, with higher power being output at port 12.
Signal S1 includes sinusoidal components at frequencies f1 and f2 having respective voltage levels of C.sub.11 V.sub.1 and C.sub.11 V.sub.2, where C.sub.11 is the coupling coefficient of coupler C1. The phases of the f1 and f2 components of signal S1 are defined to be 0.degree.. Similarly, signal S2 includes sinusoidal components at frequencies f1 and f2 having respective voltage levels V.sub.1 1+L -C.sub.11.sup.2 +L , and having respective phases also defined to be 0.degree..
Signal S1 is applied to a power amplifier A1 where it is amplified to produce an amplified signal S3 output at an amplifier output port 14. Amplifier A1 is a conventional high-frequency amplifier operating in class A, AB or B; for example, a power gain on the order of 30 dB to produce RF output power of about 50 W.
Amplified signal S3 contains amplified frequency components f1 and f2 and undesirable intermodulation distortion products at frequencies f3 and f4. Frequency f3 is at 2f1-f2, and is less than frequency f1. Frequency f4 is at 2f2-f1, and is greater than frequency f2. The components of signal S3 at frequencies f1 and f2 are G1C.sub.11 V.sub.1 and G1C.sub.11 V.sub.2, respectively, where G1 is the voltage gain of amplifier A1. The phases of the f1 and f2 components of S3 are -.phi..sub.10 and -.phi..sub.20, respectively, where -.phi..sub.10 and -.phi..sub.20 are the respective insertion phase lags through amplifier A1 at frequencies f1 and f2. The minus sign indicates a phase lag or delay. The intermodulation distortion components of signal S3 at frequencies f3 and f4 have respective voltage levels V.sub.3 and V.sub.4 with respective reference phase values of -.phi..sub.30 and -.phi..sub.40.
Signal S3 is applied to an input port 15 of a coupler C2, such as a conventional hybrid (e.g., branch line), a backward firing or a Wilkinson coupler. Coupler C2 has a coupling coefficient of C.sub.22 that is typically in the range of -10 to -20 dB. A coupled-path signal S4 is output from a coupling port 16 and is, for example, 10 to 20 dB below the level of a direct-path signal S8 that is output from a direct port 17. The voltage levels of the frequency components of signal S4 are each C.sub.22 times the corresponding voltage levels of the signal S3 frequency components. The voltage levels of the components of signal S8 are 1+L -C.sub.22.sup.2 +L times the corresponding voltage levels of the components of signal S3. The respective phases -.phi..sub.11 to -.phi..sub.41 of the frequency components f1-f4 of signal S4 are the same as the phases of the corresponding frequency components of signal S8. Specifically, phase values -.phi..sub.11 and -.phi..sub.12 are the combination of the insertion phase lags -.phi..sub.10 and -.phi..sub.20 through amplifier A1, respectively, plus the respective insertion phase lags at frequencies f1 and f2 through coupler C2. Phase values -.phi..sub.31 and -.phi..sub.41 are the insertion phase lags at the respective frequency f3 and f4 through coupler C2, plus the phase lag through amplifier A1.
Coupled-path signal S4 is applied to a phase shifter 18, such as a variable capacitor-type phase shifter, PIN diode phase shifter or a Shiffman phase shifter, for introducing a 180.degree. phase shift at each of the frequencies f1-f4. A signal S5 output from phase shifter 18 is input to a coupled port 19 of a coupler C3. Signal S5 contains the same frequency components f1-f4 at the same voltage levels as signal S4, but with the phase of each respective component shifted by 180.degree. from the corresponding components of signal S4. Specifically, the voltage levels of the f1 and f2 components of signal S5 are C.sub.22 G1C.sub.11 V.sub.1 and C.sub.22 G1C.sub.11 V.sub.2, respectively, and the respective phases are -.phi..sub.11 -180.degree. and -.phi..sub.21 -180.degree.. The voltage levels of the f3 and f4 components of signal S5 are C.sub.22 V.sub.3 and C.sub.22 V.sub.4, respectively, and the respective phases are -.phi..sub.31 -180.degree. and -.phi..sub.41 -180.degree..
Signal S2 is input to a delay line DL1, which outputs a signal S6. Signal S6 is input to a port 20 of coupler C3. Delay line DL1 introduces phase lags of -.phi..sub.11 and -.phi..sub.21 at respective frequencies f1 and f2, equalling the insertion phase lags through amplifier A1 plus coupler C2 at frequencies f1 and f2. Thus, the f1 and f2 frequency components of signal S6 are 180.degree. out-of-phase with the f1 and f2 frequency components of signal S5.
Coupler C3 substantially subtracts signal S5 from signal S6 to produce a signal S7 having signal components f1-f4. In this case, the f1 and f2 components of signal S7 have respective phase values that are equal to the phase of f1 and f2 components of signal S5 or S6, depending on the cancellation in coupler C3. The voltage levels of the f1 and f2 components of signal S7 are (1+L -C.sub.22.sup.2 +L 1+L -C.sub.22.sup.2 +L -C.sub.33 C.sub.11 C.sub.22 G1) times V.sub.1 and V.sub.2, respectively, where C.sub.33 is the coupling coefficient of coupler C3. Coupler C3 also produces the f3 and f4 components of signal S7 at voltage levels of C.sub.33 C.sub.22 V.sub.3 and C.sub.33 C.sub.22 V.sub.4, respectively, with respective phase values of -.phi..sub.32 and -.phi..sub.42.
Signal S7 is applied to the input of an amplifier A2. The respective voltage or power levels of signal S7 at frequencies f1 and f2 are below the corresponding power levels of signal S1 as a function of the cancellation in coupler C3. Amplifier A2 outputs a signal S9. The gain G2 of amplifier A2 is selected such that the f3 and f4 terms cancel at the output.
Signal S9 also contains distortion components at distortion frequencies f3 and f4, respectively designated as "f3,S9" and "f4,S9", that are primarily the result of the amplification of the corresponding distortion frequency components of signal S7. A signal S10, which is signal S8 delayed by a delay line DL2, contains distortion frequency components at respective frequencies f3 and f4 and respectively "f3,S10" and "f4,S10". Signal S10 also contains fundamental frequency components "f1,S10" and "f2,S10" at respective frequencies f1 and f2.
Output combiner 22 adds the f1 and f2 frequency components when the respective phases of f1,S9 and f2,S9 equal the corresponding phases of f1,S10 and f2,S10, and when the corresponding voltage levels of S9 and S10 are equal. Delay line DL2 equalizes the phases by adding an insertion phase lag that equals the insertion phase lag through amplifier A2 and through the path between ports 20 and 21 of coupler C3. Delay line DL2 also compensates for the insertion phases of the distortion products of signal S5 through coupler C3 and for the distortion products of signal S7 through amplifier A2 at f3 and f4.
Output combiner 22 receives signals S9 and S10 at ports 23 and 24, respectively. Combiner 22, such a 3 dB Wilkinson-type coupler, cancels the distortion frequency power within signals S9 and S10 adding the two signals 180.degree. out-of-phase. Similarly, when two equal amplitude, but in-phase components are applied to ports 23 and 24, all of the power appears at the output port 25, and is without distortion components at frequencies f3 and f4 in the ideal case.
While this conventional RF amplifier topology ideally eliminates distortion products, the conventional RF amplifier topology does not provide closed-loop control, manifesting itself by degraded performance caused by changes in component characteristics resulting from, for example, temperature variations and by component aging. What is needed is an RF amplifier topology that provides simple closed-loop control for both loops, thus eliminating performance degradation as component characteristics change with temperature variations and component aging.